DC-DC Converter

ABSTRACT

Provided is a vehicle which enables a highly-efficient DC-DC converter and a highly-efficient power supply to a load, regardless of a power supply amount of to the load. When the power supply amount to a load R 1  is a predetermined value or more, a control means  5  implements a first mode for making the switching elements S 1  to S 4  driven, and when the power supply amount of to the load R 1  is the predetermined value or less, the control means  5  implements a second mode, for making the switching elements S 3  and S 4  stopped in an OFF state, and making only the switching elements S 1  and S 2  driven.

TECHNICAL FIELD

The present invention relates to a DC-DC converter having insulatingfunction.

BACKGROUND ART

A conventionally known DC-DC converter is a device for outputting apower by converting DC power to AC power by a switching circuit,transforming this using a transformer, and converting this to DC powerby a rectifier circuit. In handling large power, it is general that afull bridge circuit is used. In this full bridge circuit, a switchingelement at the upper arm side and a switching element at the lower armside of two pairs of switching elements connected in series, are drivenalternately. That is, the switching element at the upper arm side andthe switching element at the lower arm side implement reverse on/offdrive mutually. However, it became hard switching in turn-on andturn-off of the switching element, which generated a large switchingloss, resulting in inferior efficiency.

To solve this problem, a DC-DC converter to reduce the switching lossand to improve efficiency has been disclosed in PATENT DOCUMENT 1. ThisDC-DC converter is implemented by shifting a phase of on/off drive ofone switching element connected in series and on/off drive of the otherswitching element connected in series, composing the full bridgecircuit. In this way, zero voltage switching becomes possible, andreduction of the switching loss is attained. This control system iscalled a phase shift system.

In addition, in PATENT DOCUMENT 2, there has been disclosed theattainment of enhancement of efficiency and reduction of output ripple,by making either of one set of the switches of a full bridge circuitcontinuation-on, and the other continuation-off, when a load becomeslight, in a resonant circuit.

CITATION LIST Patent Literature

Patent Literature 1: JP-A-2003-47245

Patent Literature 2: JP-A-2003-324956

SUMMARY OF INVENTION Technical Problem

The full bridge circuit of the phase shift system enables zero voltageswitching, in large power supply amount to a load, however, timerequired in charge-discharge of a parasitic capacitance of the switchingelement becomes long, due to small current flowing in a circuit, insmall power supply amount to a load. Implementing of turn-on of theswitching element in a state of this insufficient charge-discharge, asit is, provides hard switching, which provided a problem of increase inswitching loss and decrease in efficiency.

In addition, in the full bridge circuit of the phase shift system,charge-discharge of the parasitic capacitance of the switching elementis utilized, whereas the resonant circuit is based on frequency control,thus, operation principle itself differs, therefore technologyapplicable to the resonant circuit cannot be applied to a circuit of thephase shift system, even though problems to be solved are common.

It is an object of the present invention to provide a highly-efficientDC-DC converter, regardless of a power supply amount of to a load.

In addition, it is an object of the present invention to provide avehicle which enables a highly-efficient power supply to a load,regardless of a power supply amount of to a load.

Solution to Problem

To attain the above object, the DC-DC converter relevant to the presentinvention is characterized by a full bridge circuit, composed of a firstswitching leg connecting a first and a second switching elements inseries, and a second switching leg connecting a third and a fourthswitching elements in series, the first and second switching legs beingconnected in parallel, wherein DC terminals are provided between bothends of the first switching leg and between both ends of the secondswitching leg, and AC terminals are provided between the seriesconnecting point of the first and the second switching elements, and theseries connecting point of the third and the fourth switching elements;a rectifier circuit having a smoothing reactor; a first smoothingcapacitor connected to a DC power source in parallel and connected tothe DC terminals of the full bridge circuit; a second smoothingcapacitor connected to a load in parallel, and connected to a DCterminals of the rectifier circuit; a primary winding connected to theAC terminals of the full bridge circuit; a secondary winding connectedto an AC terminals of the rectifier circuit; a transformer forconnecting magnetically the primary winding and the secondary winding;and a control means for controlling the full bridge circuit, wherein thefirst, second, third and fourth switching elements are each composed ofa switch, an antiparallel diode connected to the switch in parallel, anda capacitor connected to the switch and the antiparallel diode inparallel, the DC-DC converter having a reactor component insertedbetween the AC terminals and the primary winding of the full bridgecircuit in series; when power supply amount to the load is predeterminedvalue or more, the control means implements a first mode for making thefirst, the second, the third and the fourth switching elements driven;and when power supply amount to the load is predetermined value or less,the control means implements a second mode for making one set of theswitching elements of the switching leg at one side composing the firstswitching leg and the second switching leg stopped, in an OFF state, andfor making one set of the switching elements of the switching leg at theother side composing the first switching leg and the second switchingleg driven.

In addition, a vehicle relevant to the present invention ischaracterized by mounting the DC-DC converter of the present invention.

Advantageous Effects of Invention

According to the present invention, regardless of a power supply amountof to a load, a highly-efficient DC-DC converter can be provided.

According to the present invention, regardless of a power supply amountof to a load, a vehicle, which enables a highly-efficient power supplyto a load, can be provided.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a circuit configuration diagram of the DC-DC converteraccording to Embodiment 1 of the present invention.

FIG. 2 is a drawing explaining the switching of operating mode ofEmbodiment 1.

FIG. 3 is a drawing explaining a determination method for apredetermined value Pth of Embodiment 1.

FIG. 4 is a drawing explaining the switching of operating mode based ontwo predetermined values Pth1 and 2 of Embodiment 1.

FIG. 5 is a drawing of a voltage-current waveform explaining anoperation in a light load mode M2 of Embodiment 1.

FIG. 6 is a circuit diagram explaining an operation (mode a) in a lightload mode M2 in the period (a) shown in FIG. 5.

FIG. 7 is a circuit diagram explaining an operation (mode b) in a lightload mode M2 in the period (b) shown in FIG. 5.

FIG. 8 is a circuit diagram explaining an operation (mode c) in a lightload mode M2 in the period (c) shown in FIG. 5.

FIG. 9 is a circuit diagram explaining an operation (mode d) in a lightload mode M2 in the period (d) shown in FIG. 5.

FIG. 10 is a circuit diagram explaining an operation (mode e) in a lightload mode M2 in the period (e) shown in FIG. 5.

FIG. 11 is a circuit diagram explaining an operation (mode 0 in a lightload mode M2 in the period (f) shown in FIG. 5.

FIG. 12 is a circuit diagram explaining an operation (mode g) in a lightload mode M2 in the period (g) shown in FIG. 5.

FIG. 13 is a circuit diagram explaining an operation (mode h) in a lightload mode M2 in the period (h) shown in FIG. 5.

FIG. 14 is a drawing of a voltage waveform explaining other operation ina light load mode M2 of Embodiment 1.

FIG. 15 is a circuit configuration diagram of the DC-DC converteraccording to Embodiment 2 of the present invention.

FIG. 16 is a circuit configuration diagram of the DC-DC converteraccording to Embodiment 3 of the present invention.

FIG. 17 is a schematic configuration diagram of a power source system ofa conventional electric vehicle.

FIG. 18 is a schematic configuration diagram of a power source system ofan electric vehicle according to Embodiment 4 of the present invention.

DESCRIPTION OF EMBODIMENTS

Explanation will be given in detail on embodiments of the presentinvention with reference to drawings. It should be noted that, in thefollowing explanation, voltage of the switching element in an ON-state,or voltage nearly equivalent to or lower than forward voltage drop of aantiparallel diode connected to the switching element in parallel, iscalled zero voltage, and it is called zero voltage switching or softswitching to decrease switching loss by switching ON and OFF of thisswitching element, in a state that voltage applied to the switchingelement is zero voltage.

Embodiment 1

FIG. 1 is a circuit configuration diagram of the DC-DC converter 1according to Embodiment 1 of the present invention. This DC-DC converter1 supplies power to a load R1, by transforming the voltage of a DC powersource V1. It should be noted that the DC power source V1 may besubstituted with output of other converter of a power factor correctioncircuit or the like.

In FIG. 1, the DC power source V1 and a smoothing capacitor C1 areconnected between DC terminals A-A′ of a full bridge circuit 2. Asmoothing capacitor C2 and the load R1 are connected between DCterminals B-B of a rectifier circuit 7. A primary winding NI isconnected between AC terminals C-C′ of full bridge circuit 2, and asecondary winding N2 is connected between AC terminals D-D′ of therectifier circuit 7. This primary winding N1 and secondary winding N2are magnetically connected by a transformer 6. The full bridge circuit 2is composed of a first switching leg 3 connected a first switchingelement S1 and a second switching element S2 in series, and a secondswitching leg 4 connected a third switching element S3 and a fourthswitching element S4 in series.

To switching elements S1 to S4, antiparallel diodes DS1 to DS4 areconnected, respectively. Here, in the case of using MOSFET as theseswitching elements, a body diode of MOSFET can be utilized as theantiparallel diode. In addition, the switching elements S1 to S4 haveparasitic capacitance CS1 to CS4. In this case, a snubber capacitor maybe connected to the switching elements S1 to S4 in parallel, as acapacitor. In FIG. 1, as one embodiment, MOSFET is used as the switchingelements S1 and S2, and IGBT is used as the switching elements S3 andS4. A reactor Lr is inserted in series between the AC terminals and theprimary winding N1 of the full bridge circuit 2. Here, as the reactorLr, a leak inductance of the transformer 6 may be utilized.

A rectifier circuit 7 is composed of two smoothing reactors L1 and L2and two diodes D1 and D2. At one end of the secondary winding N2, oneend of the smoothing reactor L1 and a cathode of the diode D2 areconnected, and at the other end of the secondary winding N2, one end ofthe smoothing reactor L2 and a cathode of the diode DI are connected.The other end of the smoothing reactor L1 and L2 is connected to one endof the smoothing capacitor C2, and an anode of the diode D1 and D2 isconnected to the other end of the smoothing capacitor C2. Here, theswitching element may be used instead of the diode D1 and D2. In thiscase, adoption of a synchronous rectification system enables to furtherenhance efficiency of the DC-DC converter 1.

The DC-DC converter 1 of the present invention is characterized byswitching an operating mode of the switching element in response topower supply amount to the load R1. Explanation will be given below onswitching of an operating mode with reference to FIG. 2.

FIG. 2 is a drawing explaining the switching of operating mode. Outputpower Pout is a product of output current detected by a current sensor8, and output voltage detected by a voltage sensor 9. Pth is apredetermined value set to switch the operating mode. In the case whereoutput power Pout is predetermined value Pth or more, the control means5 drives the switching elements S1 to S4, as the heavy load mode M1,which is the first mode. In this case, when the switching elements S1 toS4 are driven by the phase shift system, zero voltage switching ispossible. When output power Pout becomes predetermined value Pth orless, it is transferred to the light load mode M2, which is the secondmode. The control means 5 makes the switching elements S3 and S4 stoppedin an OFF state, and drives only the switching elements S1 and S2. Thecontrol means 5 controls output power by controlling drive frequency ofthe switching elements S1 and S2. It should be noted that, in the samedrawing, a drive signal for driving the switching elements S1 to S4 isdescribed at the lower stage of the graph, and the High side representsan ON-signal and the Low side represents an OFF-signal.

FIG. 3 is a drawing explaining a determination method for apredetermined value Pth. A Ploss-Pout straight line shown by a dottedline shows loss in each output power Pout actuated by the heavy loadmode M1, and the Ploss-Pout straight line shown by a solid line showsloss in each output power Pout actuated by the light load mode M2. Inthis way, the predetermined value Pth may be determined by size of theoutput power Pout, so that loss Ploss selects the smaller operatingmodes M1 and M2. Theoretically, by setting a cross-point of the dottedline and the solid line as Pth, the best efficiency is obtained.Naturally, the predetermined value Pth may be set arbitrarily.

Here, when output power Pout is nearly the same value as predeterminedvalue Pth, there may be frequent switching between the heavy load modeM1 and the light load mode M2. In such a case, as shown in FIG. 4, bydetermining each the predetermined value Pth1 for switching from theheavy load mode M1 to the light load mode M2, and the predeterminedvalue Pth2 for switching from the light load mode M2 to the heavy loadmode M1, there may be the case where the problem can be solved.Difference between the predetermined values Pth1 and

Pth2 may be selected and determined in view of balance betweenefficiency and switching frequency, depending on a product applying thepresent technology.

Next, explanation will be given on circuit operation in the light loadmode M2 of the DC-DC converter 1 with reference to FIG. 5 to FIG. 13.Circuit operation of the heavy load mode M1 is omitted because aconventional phase shift system can be applied. FIG. 5 is a drawing of avoltage/current waveform explaining the operation in a light load modeM2 of the DC-DC converter 1. In addition, explanation will be givenfirstly on a voltage waveform in FIG. 5. An S1 drive signal to an S4drive signal represent a drive signal waveform which the control means 5outputs to the switching elements S1 to S4, respectively. Also in thesame drawing, the switching elements S1 to S4 are made ON, when thedrive signal waveform to be output to the switching elements S1 to S4becomes High, and are made OFF, when it becomes Low. T1 voltagerepresents a waveform of voltage of the node T1 at one end side of theprimary winding N1, and T2 voltage represents a waveform of voltage ofthe node T2 at the other end side of the primary winding N1. Voltagebetween T1-T2 represents a voltage waveform where T2 voltage issubtracted from T1 voltage. Explanation will be given next on a currentwaveform in FIG. 5. Si current and S2 current represent current betweendrain-source of the switching elements S1 and S2, respectively. CS1current to CS4 current represent a current waveform flowing theparasitic capacitance CS1 to CS4, respectively. As for CS1 current toCS4 current, positive current is called charge current, and negativecurrent is called discharge current, provided that flow direction fromone end of the parasitic capacitance connected to a drain of theswitching element to the other end of the parasitic capacitanceconnected to a source of the switching element is positive. DS1 currentto DS4 current represent a current waveform flowing the antiparalleldiodes DS1 to DS4, respectively. In DS1 current to DS4 current, adirection flowing from an anode to a cathode is set to be positive, inantiparallel diodes DS1 to DS4, respectively. It should be noted thatperiods (a) to (h) partitioned by each dotted line in FIG. 5 correspondto the (mode a) to the (mode h), to be explained below, respectively. Inthe light load mode M2, drive signals of the switching elements S3 andS4 become OFF over all modes of the (mode a) to the (mode h).

(Mode a)

FIG. 6 is a circuit diagram explaining an operation (mode a) in a lightload mode M2 in the period (a) shown in FIG. 5. The switching element S1is turned-ON. Voltage between the both ends of the switching element S1is zero voltage due to conduction of the antiparallel diode DS1, andthus the switching element S1 becomes zero voltage switching. Afterthat, when current flowing the reactor Lr comes up to zero, reverserecovery current flows, which is current till reverse recovery to theantiparallel diode DS4, and current flowing the reactor Lr increases ina positive direction. After that, when the antiparallel diode DS4attains reverse recovery, current flowing the switching element S1becomes charge current of the parasitic capacitance CS4 and dischargecurrent of the parasitic capacitance CS3.

(Mode b)

FIG. 7 is a circuit diagram explaining an operation (mode b) in a lightload mode M2 in the period (b) shown in FIG. 5. Discharge of theparasitic capacitance CS3 decreases both ends voltage of theantiparallel diode DS3, and crossing the zero voltage conducts theantiparallel diode DS3. Conduction of the antiparallel diode DS3inhibits flow of discharge current of the parasitic capacitance CS3 andcharge current of the parasitic capacitance CS4. Current flowed theantiparallel diode DS3 returns to the antiparallel diode DS3 through theswitching element S1, the reactor Lr and the primary winding N1. Currentflowing this route increases gradually.

(Mode c)

FIG. 8 is a circuit diagram explaining an operation (mode c) in a lightload mode M2 in the period (c) shown in FIG. 5. The switching element S1is turned-OFF. Current flowing the antiparallel diode DS3 becomes chargecurrent to the parasitic capacitance CS1 and discharge current of theparasitic capacitance CS2. Discharge of the parasitic capacitance CS2decreases voltage of the node T1, however, voltage of the node T2maintains higher voltage than the DC voltage V1 due to conduction of theantiparallel diode DS3. In this way, voltage between the node T1 and thenode T2 enlarges in a negative direction.

(Mode d)

FIG. 9 is a circuit diagram explaining an operation (mode d) in a lightload mode M2 in the period (d) shown in FIG. 5. Discharge of theparasitic capacitance CS2 decreases both ends voltage of theantiparallel diode DS2, and crossing the zero voltage conducts theantiparallel diode DS2. Conduction of the antiparallel diode DS2inhibits flow of discharge current of the parasitic capacitance CS2 andcharge current of the parasitic capacitance CS1. Current flowed theantiparallel diode DS2 returns to the antiparallel diode DS2 through thereactor Lr and the primary winding N1. Current flowing this routeincreases gradually.

(Mode e)

FIG. 10 is a circuit diagram explaining an operation (mode e) in a lightload mode M2 in the period (e) shown in FIG. 5. The switching element S2is turned-ON. Voltage between the both ends of the switching element S2is zero voltage due to conduction of the antiparallel diode DS2, andthus the switching element S2 becomes zero voltage switching. Afterthat, when current flowing the reactor Lr comes up to zero, reverserecovery current flows, which is current till reverse recovery to theantiparallel diode DS3, and current flowing the reactor Lr increases ina negative direction. After that, when the antiparallel diode DS3attains reverse recovery, current flowing the switching element S2becomes charge current of the parasitic capacitance CS3 and dischargecurrent of the parasitic capacitance CS4. Voltage of the node T2decreases by discharge of the parasitic capacitance CS4, however,voltage of the node T1 maintains zero voltage due to conduction of theswitching element S2. In this way, voltage between the node T1 and thenode T2 approaches to zero.

(Mode f)

FIG. 11 is a circuit diagram explaining an operation (mode f) in a lightload mode M2 in the period (f) shown in FIG. 5. Discharge of theparasitic capacitance CS4 decreases both ends voltage of theantiparallel diode DS4, and crossing the zero voltage conducts theantiparallel diode DS4. Conduction of the antiparallel diode DS4inhibits flow of discharge current of the parasitic capacitance CS4 andcharge current of the parasitic capacitance CS3. Current flowed theantiparallel diode DS4 returns to the antiparallel diode DS4 through theprimary winding N1, the reactor Lr and the switching element S2. Currentflowing this route increases gradually.

(Mode g)

FIG. 12 is a circuit diagram explaining an operation (mode g) in a lightload mode M2 in the period (g) shown in FIG. 5. The switching element S2is turned-ON. Current flowing the switching element S2 becomes dischargecurrent of the parasitic capacitance CS1 and charge current of theparasitic capacitance CS2. Charge of the parasitic capacitance CS2increases voltage of the node T1, however, voltage of the node T2maintains zero voltage due to conduction of the antiparallel diode D4.In this way, voltage between the node T1 and the node T2 increases in apositive direction.

(Mode h)

FIG. 13 is a circuit diagram explaining an operation (mode h) in a lightload mode M2 in the period (h) shown in FIG. 5. Discharge of theparasitic capacitance CS1 decreases both ends voltage of theantiparallel diode DS1, and crossing the zero voltage conducts theantiparallel diode DS1. Conduction of the antiparallel diode DS1inhibits flow of discharge current of the parasitic capacitance CS1 andcharge current of the parasitic capacitance CS2. Current flowed theantiparallel diode DS1 returns to the antiparallel diode DS1 through theantiparallel diode DS4, the primary winding N1 and the reactor Lr.Current flowing this route decreases gradually.

Hereafter, by returning to the (mode a), operation of the above (mode a)to (mode h) is repeated.

It should be noted that, in the (mode a) to the (mode h), there is amode where current flowing in the smoothing reactor L1 and L2 flowsreversely, however, it can also be avoided by increasing a reactorvalue, by changing winding number ratio of the primary winding N1 andN2, or the like.

Reason for enabling control of the output power by controlling drivefrequency of the switching elements S1 and S2, is because a time periodwhen a voltage is generated between the node T1 and the node T2 ischanged. That is, by increasing the drive frequency, an effective valueof the voltage between the node T1 and the node T2 per one cycleincreases, and thus the output power can be increased. On the contrary,by decreasing the drive frequency, the output power also decreases. Toincrease the output power without increasing the drive frequency,adoption of a switching element having a large parasitic capacitance, asthe switching elements S3 and S4, is preferable because of enabling toextend the period when the voltage is generated between both ends of atransformer. In addition, a snubber capacitor may be connected to theswitching elements S3 and S4 in parallel. It is because the addition ofcharge-discharge period of the snubber capacitor in the (mode a) and the(mode e) results in extending the period when the voltage is generatedbetween the node T1 and the node T2. As other method for increasing theoutput power without increasing the drive frequency, a diode having slowreverse recovery characteristics can be adopted as the antiparalleldiodes DS3 and dS4. In the (mode d) and the (mode h), the voltagebetween the node T1 and the node T2 is maintained till completion ofreverse recovery of the antiparallel diodes DS3 and DS4. Accordingly,the output voltage can be increased.

Use of a switching element having fast switching characteristics as theswitching elements S1 and S2 may sometimes enhance efficiency.Generally, a MOSFET has fast switching characteristics and smallswitching loss. In addition, an LGBT has small on-resistance and smallconduction loss. For example, the MOSFET is used as the switchingelements S1 and S2, and the IGBT is used as the switching elements S3and S4. In this way, switching loss in the light load mode M2 can alsobe decreased while suppressing conduction loss in the heavy load modeM1.

On the contrary, use of the IGBT as the switching elements S1 and S2,and the MOSFET as the switching elements S3 and S4 enables to increasethe output power. Generally, a body diode of MOSFET has slow reverserecovery characteristics. Utilization of the body diode of MOSFET to theantiparallel diodes DS3 and DS4 enables to maintain voltage between thenode T1 and the node T2 till completion of reverse recovery of theantiparallel diodes DS3 and DS4, in the (mode e) and the (mode a).Accordingly, output voltage can be increased. It should be noted that itis clear that, also in the case of using MOSFET as the switchingelements S1 and S2, and IGBT as the switching elements S3 and S4, asshown in FIG. 14, by making the switching elements S1 and S2 stopped inan OFF state, and by making only the switching elements S3 and S4driven, similar effect to in the DC-DC converter having a configurationusing IGBT as the switching elements S1 and S2, and MOSFET as theswitching elements S3 and S4, can be obtained. In addition, on thecontrary, also in the case of using IGBT as the switching elements S1and S2, and MOSFET as the switching elements S3 and S4, as shown in FIG.14, by making the switching elements S1 and S2 stopped in an OFF state,and by making only the switching elements S3 and S4 driven, it is clearthat similar effect to in the DC-DC converter having a configurationusing MOSFET as the switching elements S1 and S2, and IGBT as theswitching elements S3 and S4, can be obtained.

As described above, the DC-DC converter 1 of the present invention ischaracterized in that, zero voltage switching becomes easy to berealized, even in the light load. However, when power supply amount tothe load is considered to be nearly zero, a current required forcharge-discharge of the parasitic capacitance CS 1 to CS4 cannot besecured, which may sometimes provide a case that the switching elementsS1 and S2 become hard switching. However, in this case, a drivefrequency of the switching elements S1 and S2 is lower as compared witha drive frequency in the heavy load mode M1. Therefore, it can be saidthat the present invention is effective, because even when power supplyamount to the load appears nearly zero, the light load mode M2 hashigher efficiency than the heavy load mode M1.

In addition, in the above PATENT LITERATURE 2, a resonant type circuithaving operation principle different from that of the phase shift typeis shown. In the resonant type circuit, the frequency range should belimited for stable operation. Also, many limitations in input voltagerange and variation range of output voltage. In addition, it requireshaving operation frequency apart from resonance frequency to turn downthe output because the resonant circuit is a frequency controlledcircuit. Therefore, it causes increase in ripple, requires bigger powerto drive the elements, and thus making higher efficiency is difficult toattain.

On the contrary, in a circuit of the phase shift type, the operation isimplemented by utilization of conduction of a diode connected to theseswitching elements in parallel or charge-discharge to the parasiticcapacitance, in addition to ON/FF of the switching element. And, inorder to improve efficiency, it is important how to realize zero voltageswitching or switching near thereto, in performing ON/FF of theswitching element. Therefore, it is important that the charge-dischargeto the parasitic capacitance is controlled.

Conventionally, in the light load, because sufficient current does notflow in a circuit, output capacity of a switch is not charged-dischargedsufficiently, providing hard switching, thus incurring deterioration ofefficiency. However, in the present Embodiment, this problem iseliminated by making operation of one switching legs composed of one setof switching elements connected in series in the full bridge circuitstopped in the case of light load. After confirming a current stateflowing in the circuit in such a state, it was confirmed that thecurrent for charge-discharge of the output capacity of a switch in thelight load has increased, as compared with conventional control,although the reason for that has not yet been clarified. In this way,charge-discharge of the output capacity of a switch is prompted, even inthe light load, making soft switching possible.

That is, according to the present Embodiment, the turn-on of the switchbecomes possible by lower voltage, and switching loss decreases, ascompared with a conventional control method. In addition, according tothe present Embodiment, the output is more easily turned down due to lowfrequency, as well as because of making operation of one set of theswitching elements connected in series composing one switching legsstopped, drive loss in theses switches can also be suppressed, and thusfurther enhancement of efficiency can be attained, as compared with aconventional control method.

Embodiment 2

FIG. 15 is a circuit configuration diagram of the DC-DC converter 101according to Embodiment 2 of the present invention. The same referencesigns will be furnished to the same parts as in FIG. 1, and explanationthereof is omitted. The rectifier circuit 7 is composed of the smoothingreactor L11 and the two diodes D1 and D2. One end of the smoothingreactor L11 is connected to the diodes D1 and D2, and the other end ofthe smoothing reactor L11 is connected to one end of the smoothingcapacitor C2. Two secondary winding N21 and N22 are connected mutuallyat one end thereof, and the connecting point thereof is connected to theother end of the smoothing capacitor C2. As for other end of thesecondary winding N21 and N22, N21 and N22 are connected to the anode ofthe diode D1 and the anode of the diode D2, respectively. In this way,the smoothing reactor can be reduced, by which parts number can bereduced and cost can be decreased, as compared with Embodiment 1. Inaddition, by using the switching element instead of the diodes D1 andD2, and by using a synchronous rectifying system, further higherefficiency can be attained.

Embodiment 3

FIG. 16 is a circuit configuration diagram of the DC-DC converter 102according to Embodiment 3 of the present invention. The same referencesigns will be furnished to the same parts as in FIG. 1, and explanationthereof is omitted. The rectifier circuit 7 is composed of the smoothingreactor L12, the diode leg 10 connected the diodes D1 and D2 in series,and the diode leg 11 connected the diodes D3 and D4 in series andconnected to the diode leg 10 in parallel. One end of the smoothingreactor L12 is connected to one end of the diode leg 10, the other endof the smoothing reactor L12 is connected to one end of the smoothingcapacitor C2, and the other end of the diode leg 10 is connected to theother end of the smoothing capacitor C2. The connecting point the diodesD1 and D2, and the connecting point the diodes D3 and D4 are connectedto both ends of the secondary winding N2. In this way, a diode havingsmall peak inverse voltage can be used. Such a configuration is suitableto be used when output voltage is large. In addition, by using theswitching element instead of the diodes D1 to D4, and by using asynchronous rectifying system, further higher efficiency can beattained.

Embodiment 4

FIG. 17 is a schematic configuration diagram of a power source system ofa conventional electric vehicle 31. The charger 32 converts AC powerfrom the AC power source 51 to DC power with the AC-DC converter 52, andthe DC-DC converter 53 supplies a power, by transforming DC power tovoltage required for charging the battery 41. On the other hand, theDC-DC converter 55 supplies power to the load 56, by transformingvoltage of the battery 42, which is lower voltage than voltage of thebattery 41. In large power supply amount to the load 56, power of thebattery 41 is supplied to the DC-DC converter S55 and the battery 42 bythe DC-DC converter 54. However, in the case of small power supplyamount to the load 56, such as in charging the battery 41 from the ACpower source 51, there was a problem of decrease in power conversionefficiency of the DC-DC converter 54. Accordingly, the charger 32 hadthe DC-DC converter 57 to supply power from the AC-DC converter 52 tothe DC-DC converter 55 and the battery 42 from the DC-DC converter 57not via the DC-DC converter 54.

FIG. 18 is a schematic configuration diagram of a power source system ofan electric vehicle 131 adopting the DC-DC converter 1 according toEmbodiment 4 of the present invention. The same reference code will befurnished to the same parts as in FIGS. 17, and explanation thereof isomitted. Adoption of the DC-DC converter 1 explained in the aboveEmbodiment 1, instead of the DC-DC converter 54 in FIG. 17, enables forthe DC-DC converter 1 to supply power in high efficiency, even in thecase of small power supply amount to the load 56. In this way, thecharger 132 requires no DC-DC converter 57 in FIG. 17, enables to reduceparts number and to supply power in high efficiency, while attainingsignificant cost down.

During charging of the battery 41 by the charger 132 from the AC powersource 51, in the vehicle 131, drive frequency of the switching elementsS1 and S2 of the DC-DC converter 1 becomes lower in many cases, ascompared with drive frequency in the heavy load mode M1. That is, at atime when a vehicle itself is not used such as at midnight, the battery41 is in a charging state. In such a time, the load 56 is in a verysmall level minimum required. Therefore, even when power supply amountto a load is considered to be nearly zero, the light load mode M2 hashigher efficiency than the heavy load mode M1, and thus it can be saidthat utilization of the DC-DC converter explained in the presentEmbodiment, for an electric vehicle is very useful. It should be notedthat explanation was given in the present Embodiment on an applicationembodiment of applying the DC-DC converter explained in Embodiment 1 tothe vehicle 131, however, application of the DC-DC converter explainedin Embodiment 2 or Embodiment 3 to the vehicle 131 is similarlyeffective.

REFERENCE SIGNS LIST

1, 53 to 55, 57, 101, 102 DC-DC converter

2 full bridge circuit

3, 4 switching leg

5 control means

6 transformer

7 rectifier circuit

8 current sensor

9 voltage sensor

10, 11 diode leg

31 electric vehicle

32 charger

41,42 battery

51 AC power source

52 AC-DC converter

56,R1 load

131 electric vehicle

132 charger

V1 DC power source

C1,C2 smoothing capacitor

L1, L2, L11 ,L12 smoothing reactor

Lr reactor

N1, N2 winding

S1˜S4 switching element

DS1˜DS4 antiparallel diode

CS1˜CS4 parasitic capacitance

M1 heavy load mode

M2 light load mode

Pout output power

Pth, Pth1, Pth2 predetermined value

D1˜D4 diode

T1, T2 node

1. A DC-DC converter, comprising: a full bridge circuit, composed of afirst switching leg connecting a first and a second switching elementsin series, and a second switching leg connecting a third and a fourthswitching elements in series, the first and second switching legs beingconnected in parallel, wherein DC terminals are provided between bothends of the first switching leg and between both ends of the secondswitching leg, and AC terminals are provided between the seriesconnecting point of the first and the second switching elements, and theseries connecting point of the third and the fourth switching elements;a rectifier circuit having a smoothing reactor; a first smoothingcapacitor connected to a DC power source in parallel and connected tothe DC terminals of the full bridge circuit; a second smoothingcapacitor connected to a load in parallel, and connected to a DCterminals of the rectifier circuit; a primary winding connected to theAC terminals of the full bridge circuit; a secondary winding connectedto an AC terminals of the rectifier circuit; a transformer forconnecting magnetically the primary winding and the secondary winding;and a control means for controlling the full bridge circuit, wherein thefirst, second, third and fourth switching elements are each composed ofa switch, an antiparallel diode connected to the switch in parallel, anda capacitor connected to the switch and the antiparallel diode inparallel, the DC-DC converter having a reactor component insertedbetween the AC terminals and the primary winding of the full bridgecircuit in series; when power supply amount to the load is predeterminedvalue or more, the control means implements a first mode for making thefirst, the second, the third and the fourth switching elements driven;and when power supply amount to the load is predetermined value or less,the control means implements a second mode for making one set of theswitching elements of the switching leg at one side composing the firstswitching leg and the second switching leg stopped, in an OFF state, andfor making one set of the switching elements of the switching leg at theother side composing the first switching leg and the second switchingleg driven.
 2. The DC-DC converter according to claim 1, wherein: inimplementing the first mode, the control means drives the first, thesecond, the third and the fourth switching elements in a phase shiftsystem; and in implementing the second mode, the control means drivesone set of the switching elements of the switching leg at the side todrive, in a frequency control system.
 3. The DC-DC converter accordingto claim 1, wherein the capacitor, which one set of the switchingelements of the switching leg at the side to drive in implementing thesecond mode has, has a capacitance larger than that of the capacitorwhich one set of switching elements of the switching leg at the side tobe stopped in implementing the second mode has.
 4. The DC-DC converteraccording to claim 1, wherein a snubber capacitor is connected to eachof one set of switching elements of the switching leg at the sidestopped in implementing the second mode in parallel.
 5. The DC-DCconverter according to claim 1, wherein the antiparallel diode, whichone set of the switching elements of the switching leg at the sidestopped in implementing the second mode has, has a reverse recoverycharacteristic slower than that of the antiparallel diode which one setof the switching elements of the switching leg at the side driven inimplementing the second mode has.
 6. The DC-DC converter according toclaim 1, wherein one set of the switching elements of the switching legat the side to drive in implementing the second mode has a switchingcharacteristic faster than that of one set of the switching elements ofthe switching leg at the side to be stopped in implementing the secondmode.
 7. The DC-DC converter according to claim 1, wherein one set ofthe switching elements of the switching leg at the side to drive inimplementing the second mode are MOSFETs, and one set of the switchingelements of the switching leg at the side to be stopped in implementingthe second mode are IGBTs.
 8. The DC-DC converter according to claim 1,wherein one set of the switching elements of the switching leg at theside to drive in implementing the second mode are IGBTs, and one set ofthe switching elements of the switching leg at the side to be stopped inimplementing the second mode are MOSFETs.
 9. The DC-DC converteraccording to claim 1, wherein the predetermined value includes a firstpredetermined value and a second predetermined value larger than thefirst predetermined value, and the control means is configured to switchto a second mode, in the case where a power supply amount to the load isthe first predetermined value or less, and is configured to switch to afirst mode, in the case where power supply amount to the load is thesecond predetermined value or more.
 10. The DC-DC converter according toclaim 1, wherein the rectifier circuit is provided with a connectingbody between one end of the first smoothing reactor and one end of thesecond smoothing reactor, and a connecting body between one end of thefirst diode and one end of the second diode, the other end of the firstsmoothing reactor being connected to the other end of the first diode,and the other end of the second smoothing reactor being connected to theother end of the second diode, wherein AC terminals are provided betweenthe other end of the first diode and the other end of the second diode,and DC terminals are provided between the connecting point of the firstand the second smoothing reactors and the connecting point of the firstand the second diodes.
 11. The DC-DC converter according to claim 1,wherein the rectifier circuit is provided with a connecting body betweenone end of the first smoothing reactor and one end of the secondsmoothing reactor, and a connecting body between one end of theswitching element at the first rectifier circuit side and one end of theswitching element at the second rectifier circuit side, the other end ofthe first smoothing reactor being connected to the other end of theswitching element at the first rectifier circuit side, and the other endof the second smoothing reactor being connected to the other end of theswitching element at the second rectifier circuit side, wherein ACterminals are provided between the other end of the switching element atthe first rectifier circuit side and the other end of the switchingelement at the second rectifier circuit side, and DC terminals areprovided between the connecting point of the first and the secondsmoothing reactors and the connecting point of the switching elements atthe first and the second rectifier circuit sides.
 12. The DC-DCconverter according to claim 1, wherein the secondary winding isprovided with a connecting body between one end of the first secondarywinding and one end of the second secondary winding, and the rectifiercircuit is provided with the smoothing reactor and the first and thesecond diodes, one end of the first diode being connected to the otherend of the first secondary winding, one end of the second diode beingconnected to the other end of the second secondary winding; and theother end of the first diode and the other end of the second diode beingconnected to one end of the smoothing reactor, wherein DC terminals areprovided between the connecting point of the first and the secondsecondary winding and the other end of the smoothing reactor, and ACterminals are provided between one end of the first diode and one end ofthe second diode.
 13. The DC-DC converter according to claim 1, whereinthe secondary winding is provided with a connecting body between one endof the first secondary winding and one end of the second secondarywinding, and the rectifier circuit is provided with the smoothingreactor and the switching elements at the first and the second rectifiercircuits sides, one end of the switching element at the first rectifiercircuit side being connected to the other end of the first secondarywinding, one end of the switching element at the second rectifiercircuit side being connected to the other end of the second secondarywinding, the other end of the switching element at the first rectifiercircuit side and the other end of the switching element at the secondrectifier circuit side being connected to one end of the smoothingreactor, wherein DC terminals are provided between the connecting pointof the first and the second secondary windings and the other end of thesmoothing reactor, and AC terminals are provided between one end of theswitching element at the first rectifier circuit side and one end of theswitching element at the second rectifier circuit side.
 14. The DC-DCconverter according to claim 1, wherein the rectifier circuit includes afirst diode leg connecting a smoothing reactor and a first and a seconddiodes in series, and a second diode leg connecting a third and a fourthdiodes in series and connected to the first diode leg in parallel, oneend of the smoothing reactor being connected to one end of the firstdiode leg, wherein DC terminals are provided between the other end ofthe smoothing reactor and the other end of the first diode leg, and ACterminals are provided between the series connecting point of the firstand the second diodes and the series connecting point of the third andthe fourth diodes.
 15. The DC-DC converter according to claim 1, whereinthe rectifier circuit includes a first switching leg at the rectifiercircuit side connecting a smoothing reactor a first switching elementand a second switching element at the rectifier circuits sides inseries, and a second switching leg at the rectifier circuit sideconnecting a third switching element and a fourth switching element atthe rectifier circuits sided in series and connected to the firstswitching leg at the rectifier circuit side in parallel, one end of thesmoothing reactor being connected to one end of the first switching legat the rectifier circuit side, wherein DC terminals are provided betweenthe other end of the smoothing reactor and the other end of theswitching leg at the first rectifier circuit side, and AC terminals areprovided between the series connecting point of the first and the secondswitching elements at the rectifier circuits side, and the seriesconnecting points of the third and the fourth switching elements at therectifier circuit side.
 16. (canceled)
 17. (canceled)